A conventional digital radiography (DR) imaging panel acquires image data from a scintillating medium using an array of individual sensors, arranged in a row-by-column matrix, in which each sensor provides a single pixel of image data. Each pixel generally includes a photosensor and a switching element which can be arranged in a planar or a vertical manner, as is generally known in the art. In one known imaging arrangement, a frontplane has an array of photosensitive elements, and a backplane consists of an array of thin-film transistor (TFT) switches. In these imaging devices, hydrogenated amorphous silicon (a-Si:H) is commonly used to form the photodiode and the thin-film transistor switch needed for each pixel, although polycrystalline semiconductors such as laser recrystallized silicon and single-crystal silicon TFT switches can alternately be employed.
FIG. 1 shows a schematic of a portion of a flat panel imager 80 of this conventional type consisting of an array having a number of a-Si:H n-i-p photodiodes 70 and TFTs 71. Gate driver chips 82 connect to the blocks of gate lines 83 and readout chips 34 connect to blocks of data lines 84 and bias lines 85. Charge amplifiers 86 may be provided that receive signals from the data lines. An output from the charge amplifiers 86 may go to an analog multiplexer 87 or directly to an analog-to-digital converter (ADC) 88 to stream out the digital image data at desired rates.
In the conventional a-Si:H-based indirect flat panel imager of FIG. 1, incident X-ray photons are converted to optical photons, which are subsequently converted to electron-hole pairs within the a-Si:H n-i-p photodiodes 70. The pixel charge capacity of the photodiodes is a product of the bias voltage and the photodiode capacitance. In general, a reverse bias voltage is applied to the bias lines 85 to create an electric field (and hence a depletion region) across the photodiodes and enhance charge collection efficiency. The image signal is integrated by the photodiodes while the associated TFTs 71 are held in a non-conducting (“off”) state. This is accomplished by maintaining the gate lines 83 at a negative voltage. The array is read out by sequentially switching rows of the TFTs 71 to a conducting state by means of TFT gate control circuitry. When a row of pixels is switched to a conducting (“on”) state by applying a positive voltage to the corresponding gate line 83, charge from those pixels is transferred along data lines 84 and integrated by external charge-sensitive amplifiers 86. The row is then switched back to a non-conducting state, and the process is repeated for each row until the entire array has been read out. The signal outputs from the external charge-sensitive amplifiers 86 are transferred to an analog-to-digital converter (ADC) 88 by a parallel-to-serial multiplexer 87, subsequently yielding a digital image.
The flat panel imager having an imaging array as described with reference to FIG. 1 is capable of both single-shot (radiographic) and continuous (fluoroscopic) image acquisition. However, the charge amplifiers of the conventional circuit arrangement are subject to common mode noise and other problems that constrain signal quality.
Also known in the art are digital radiographic imaging panels that utilize an array of pixels comprising an X-ray absorbing photoconductor, such as amorphous Selenium (a-Se), and a readout circuit. Since the X-rays are absorbed in the photoconductor, no separate scintillating screen is required.
These conventional imaging arrays have limitations which affect performance. One limitation of digital radiographic arrays employing MIS photosensors, for example, is a reduction in quantum efficiency due to capacitive division of charge between the capacitance of the gate dielectric and capacitance of the semiconductor. Conventional MIS photosensor architectures are well known to those of ordinary skill in the art, and thus will not be described herein in detail. For convenience, an equivalent circuit of an MIS photosensor is shown in FIG. 2A. That circuit consists of the capacitance of the insulator, given byCi=εiti,in which ∈i is the dielectric constant of the insulator and ti is the insulator thickness, in series with the capacitance of the semiconductorCs=εsts,in which εs is the dielectric constant of the semiconductor and ts is the semiconductor thickness. The thermal generation of charge in the semiconductor and the photo-generation of charge in the semiconductor act as current sources in parallel with the semiconductor capacitance. In the operation of the MIS photosensor, a reverse bias is applied between the common bias electrode contacting the semiconductor and the pixel electrode contacting the insulator. When the semiconductor of the MIS photosensor consists of an N-type doped layer overlying an intrinsic semiconductor layer, the semiconductor contact would be positive biased and the electrode contacting the insulator would be negative biased.
Exposure to light with energy levels above the semiconductor band gap results in creation of one electron-hole pair per absorbed photon and generates a charge difference between photosensor terminals. Due to the electric field in the semiconductor resulting from the applied bias, the electrons flow out through the N+ semiconductor into the bias line while the holes drift to the interface between the semiconductor and the insulator. The mirror charge on the electrode contacting the gate dielectric is less than the number of holes at the silicon-insulator interface due to capacitive division between the insulator capacitance Ci and the semiconductor capacitance CS. The photocurrent, IPHOTO in FIG. 2A, results in charge separation across the insulator capacitance Ci with holes at the interface between the insulator and semiconductor and electrons in the top electrode. For convenience, this reduction in charge is called the charge transfer efficiency, or CTE:CTE=Ci/(Ci+Cs).The quantum efficiency, defined as the number of electrons collected on the electrode contacting the insulator divided by the number of incident photons, is reduced by the CTE:QE=TITO·exp(−αλdn-layer)·(1−exp(−αλdi-layer)·CE·CTEWhere TITO is the optical transmission of the transparent electrode (typically Indium Tin Oxide), αλ is the optical absorption coefficient of the semiconductor (typically amorphous silicon) and dn-layer and di-layer are the thicknesses of the doped layer (typically n-doped amorphous silicon) and the intrinsic layer, respectively, and CE is the charge collection efficiency in the intrinsic layer.
FIG. 2B shows the charge transfer efficiency as a function of the semiconductor and insulator thicknesses for the particular case of amorphous silicon semiconductor and silicon nitride insulator. For values typically used in prior art digital radiographic detectors employing MIS photosensors, the charge transfer efficiency can be as low as 50%-75%. This is a problem with the prior art devices.
In addition to low quantum efficiency, prior art devices also are subject to noise sources, including at least common-mode noise from gate line switching, power supply noise and ripple, and electromagnetic interference (EMI) pickup.
FIG. 3A shows a circuit diagram for a prior-art, thin-film transistor array with a 1-transistor passive pixel architecture with MIS photosensors and FIG. 3B shows a circuit diagram for a prior-art thin-film transistor array with a 1-transistor passive pixel architecture with PIN photosensors. In operation of these two architectures, the pixel is first reset by establishing a reverse bias across the photosensor. This is accomplished by closing a charge amplifier reset switch and turning the row select TFTs on. The anode of the photosensor is then set to Vbias, while the cathode of the photosensor is set to Vref. For the reverse bias condition Vref>VBIAS. Once the pixel is reset, the row select TFT is turned off and the device is isolated. Upon signal detection, electron hole pairs are generated, providing a charge difference between photosensor terminals, and are swept to the contacts by the electric field. These carriers remove existing charge at the terminals, effectively reducing the bias across the diode. During the read out stage, the amount of positive charge that is required to re-establish the initial reverse bias condition is measured through the use of the charge amplifier. This analog output signal that corresponds to this charge difference is then either converted directly or indirectly to provide digital values, depending on the signal processing techniques employed.
FIG. 3C illustrates the capacitive coupling between the gate lines and the data lines in the circuits of FIGS. 3A and 3B. These overlap capacitances include the physical overlaps of gate line and data line as well as the capacitances between the sources of the thin-film row select transistors and the gates of those transistors. Any noise or ripple on the gate line clocks feeds through onto the data lines in the ratio of the sum of the gate line to data line overlap capacitances to the total capacitance of the data line:ΔVnoise feedthru=ΔVnoise·Nrows·Coverlap/Cdataline, where ΔVfeedthru is the voltage noise appearing on the data lines due to a ΔVnoise noise voltage on the gate lines, Nrows is the number of rows in the image sensor, Coverlap is the overlap capacitance between one row select line and one data line, and Cdataline is the total data line capacitance. The corresponding noise charge ΔQnoise feedthru sensed by the charge amplifier is given by:ΔQnoise feedthru=ΔVnoise·Nrows·Coverlap. 
For a typical prior-art image sensor, the total overlap capacitance Nrows·Coverlap is typically half or more of the data line capacitance. For a typical 10 mV noise voltage on the gatelines, for example from power supply noise or clock driver noise, the noise voltage on the data line would be 5 mV, which is well within the signal range for the diagnostic region of interest for digital radiographic applications such as chest radiography, mammography, or fluoroscopy.
In most prior-art thin-film transistor arrays for digital radiographic sensing, this ratio of overlap capacitance to total data line capacitance is typically 0.5 to 0.9 since the overlap capacitance comprises the majority of the data line capacitance. Sources of noise or ripple in the gate line power supplies include noise from switching in switching power supplies, electromagnetic pick-up on power supply bias lines, and circuit noise in the integrated circuit used for generating the row-select clock pulses. Since this noise is usually temporally uncorrelated with the array readout timing, it cannot be removed by calibration.
A second source of noise in prior-art thin-film transistor arrays for digital radiography is the feed-through of the gate line row-select clock onto the data line. During readout, row select is performed by clocking the gate lines sequentially between an “off” voltage which maintains the TFT switch in the pixel in a high-resistance state to an “on” voltage which turns the TFT switch in the pixels in that row to a low-resistance state. For amorphous silicon or polycrystalline silicon thin film transistors, this voltage is typically 20V or higher. The feedthrough voltage can be approximated by:Vfeedthru=ΔVrow select·Coverlap/Cdataline And the feedthrough charge Qfeedthru is approximated byQfeedthru=ΔVrow select·Coverlap Where ΔVrow select is the voltage swing on the row select line, Coverlap is the overlap capacitance between a row select line (gate line) and a data line, and Cdataline is the total data line capacitance. Since the overlap capacitance is generally half or more of total data line capacitance in thin-film TFT arrays for digital radiographic applications, the resulting feedthrough voltage can be significant as compared to the signal charge. For a typical prior-art radiographic array with 2,000 rows, the feedthrough voltage from a 20V clock pulse would be ˜10 mV, equivalent to a signal level well within the diagnostic region of interest for most radiographic applications. This coupling is not entirely reversed when the device is subsequently switched off, making it potentially difficult to calibrate for such an offset.
A third source of noise in prior-art thin-film transistor arrays for digital radiography is the electromagnetic pick-up on the data lines. In radiographic imaging applications, the array dimensions range from 12 inches to 17 inches in length, the 12-inch to 17-inch long data lines act as antennas that pick up signals from stray electromagnetic fields. These stray electromagnetic fields are caused by sources such as electromagnetic emission from circuit boards supporting the radiographic imaging array, which are normally in close proximity to the array, electromagnetic fields from the X-ray generator used in conjunction with the radiographic imaging panel, electromagnetic fields from building power lines, radio-frequency communications, equipment operating in the proximity of the radiographic imaging system, and other sources of electromagnetic fields external to the panel.
Given these difficulties, it is apparent that there would be advantages to readout circuitry solutions that minimize or eliminate noise-related problems, particularly for systems using lower radiation levels.
A second class of prior-art thin-film-transistor image sensing arrays, termed active pixel imaging arrays, utilize amplifiers within each pixel. Whether utilizing MIS photosensors, PIN photodiodes, or other photosensors, these active pixel imaging arrays also suffer from the same difficulties as described above for the passive pixel arrays.
Thus, existing devices exhibit relatively low overall quantum efficiency and are subject to noise sources, including common-mode noise from gate line switching, power supply noise and ripple, and electromagnetic interference (EMI) pickup. Existing panels do not achieve limited quantum performance, even at radiation levels as high as 1.0 mR. There can be a large offset (for example, an offset of 200 mV or more) in signal noise due to coupling through channel capacitance when an address switching TFT is switched off. This coupling is not reversed when the device is subsequently switched on, making it potentially difficult to calibrate for such an offset. Long data lines, some in excess of 17 inches, characteristically form antenna structures for EMI. This is not easily compensated by grounding techniques. The addition of EMI shielding, while it has some beneficial effects, is costly and adds weight. Gate line voltage noise feeds through to data lines due to trace overlap capacitance, so that TFT gate-to-drain capacitance effectively appears on the data lines.
Thus, given these difficulties, it is apparent that there would be advantages to readout circuitry solutions that minimize or eliminate noise-related problems, particularly for systems using relatively lower radiation levels.